1. Field of the Invention
The prevent invention relates to a converter having a power factor correcting function.
2. Description of Related Art
FIG. 1 is a circuit diagram illustrating a converter according to a related art. In FIG. 1, the converter includes a PFC (power factor correction) part A to correct a power factor and a DC-DC converter part B that employs a transformer to realize electric insulation and convert a voltage into another voltage. The PFC part A rectifies an AC voltage supplied from an AC power source AC, carries out ON/OFF control of a switching element Qp with a step-up chopper so that an input current may have the same waveform as an input voltage, and controls an output voltage Vp of the PFC part A to a constant value. The DC-DC converter part B converts, in an insulated manner, the output voltage Vp from the PFC part A into an optional output voltage Vo. The DC-DC converter part B is, for example, a half-wave-rectifying current resonance converter.
Operation of the converter of FIG. 1 will be explained. A sinusoidal voltage from the AC power source AC is rectified by a bridge rectifier DB through a filter circuit FL, to provide the step-up chopper with a full-wave-rectified waveform. The step-up chopper includes a winding N1 of a reactor L1 having a transformer configuration, the switching element Qp of a MOSFET, and a rectifying diode Dp.
First, a flip-flop FF is set, a gate waveform (signal) illustrated in FIG. 2 is applied to a gate of the switching element Qp to turn on the switching element Qp, and a current passes clockwise through a path extending along AC, FL, DE, N1 of L1, Qp, R5, DB, FL, and AC, to accumulate energy in the reactor L1. A switching current passing through the switching element Qp is detected by the resistor R5 for detecting switching current as a voltage VR5 illustrated in FIG. 2 and is compared by a comparator COMP2 with a target value VM.
If the switching current reaches the target value VM, the flip-flop FF is reset to turn off the switching element Qp. When the switching element Qp is turned off, a combination of the energy accumulated in the reactor L1 and the voltage supplied from the AC power source AC charges through the rectifying diode Dp an output capacitor Ci of the PFC part A. The voltage supplied to the output capacitor Ci of the PFC part A is higher than a peak value of the supplied sinusoidal voltage. The voltage Vp of the output capacitor Ci of the PFC part A is detected by resistors R6 and R7 and is compared by an operational amplifier OTA with a second reference voltage ES2. An error signal between the voltage Vp detected through the resistors R6 and R7 and the second reference voltage ES2 is supplied from the operational amplifier OTA to a multiplier MUL. The multiplier MUL multiplies the full-wave-rectified waveform detected through resistors R1 and R2 by the error signal and supplies a result of the multiplication to the comparator COMP2 as the switching current target value VM.
After the reactor L1 discharges the energy, a voltage VN2 of a criticality detecting winding N2 is inverted as illustrated in FIG. 2. A comparator COMP1 compares the voltage VN2 with a first reference voltage ES1 and sets the flip-flop FE. This again turns on the switching element Qp. Thereafter, the above-mentioned operation is repeated to generate a control signal for the switching element Qp so that the voltage Vp of the output capacitor Ci of the PFC part A may keep a constant value and so that an input current may have a sinusoidal waveform that follows an input voltage waveform. The voltage Vp of the output capacitor Ci of the PFC part A serves as a DC power source for the DC-DC converter part B.
FIG. 3 illustrates an example of a controller 1 arranged in the DC-DC converter part B and FIG. 4 illustrates waveforms at various locations of the controller 1. Operation of the controller 1 will be explained.
In FIG. 3, an oscillator OSC outputs a pulse (OSC output in FIG. 4) to a one-shot circuit OST of a one shot multivibrator. Based on the pulse from the oscillator OSC, the one-shot circuit OST outputs a pulse (OST output in FIG. 4) having a predetermined pulse width to a dead time generator DT1. When the pulse rises, a dead time is added (DT1 output in FIG. 4).
At the same time, the output of the one-shot circuit OST is inverted by an inverter INV and the inverted output (INV output in FIG. 4) is supplied to a dead time generator DT2. The output of the dead time generator DT1 is supplied to a buffer BUF1, which provides a drive signal LD to drive a low-side switching element Q1. An output (DT2 output in FIG. 4) from the dead time generator DT2 is changed by a level shifter LES into a signal having a different potential level and the level shifted signal is given to a buffer BUF2. The buffer BUF2 outputs a drive signal HD to drive a high-side switching element Q2. The oscillation frequency of the oscillator OSC is controlled according to a current provided by a feedback terminal FB. As the feedback terminal current IFB increases, the oscillation frequency increases.
The switching elements Q1 and Q2 each are made of a MOSFET, each have a given dead time, and are alternately turned on/off. FIGS. 5A and 5B illustrate waveforms in the DC-DC converter part B.
When the switching element Q2 turns on, a current IQ2 passes clockwise through a path extending along C1, Q2, Lr, P, Cri, and Ci. At this time, the waveform of the current IQ2 is mostly determined by a resonance frequency of the current resonance capacitor Cri and “Lr+Lp”. Here, Lp is an inductance of the primary winding P of a transformer Ta. The resonance frequency at this time is sufficiently lower than the switching frequency of the switching element Q2 and a sinusoidal wave is partly observed as a triangular wave (refer to IQ2 in FIG. 5A). The current IQ2 serves as an excitation current for the primary winding P of the transformer Ta.
When the switching element Q2 is turned off while the current IQ2 is passing, a voltage across the switching elements Q1 and Q2 has a quasi-voltage-resonance waveform based on a resultant value of the voltage resonance capacitor Crv and current resonance capacitor Cri and a resultant value of the inductance Lp of the primary winding P of the transformer Ta and the leakage inductance Lr. According to a relationship of Crv<<Cri, a resonance frequency at this time is mostly determined by the voltage resonance capacitor Crv.
The excitation current for the primary winding P passing through the switching element Q2 is translocated to a parasitic diode of the switching element Q1. After a voltage VQ1 across the switching element Q1 reaches zero, the switching element Q1 is turned on, to realize zero-voltage switching. Thereafter, a current IQ1 translocated to the switching element Q1 decreases, inverts its polarity, and passes through a MOSFET part of the switching element Q1. As a result, the current IQ1 passes counterclockwise through a path extending along Cri, P, Lr, Q1, and Cri. A current waveform at this time is of a resultant current of a resonance current created by the current resonance capacitor Cri having a highest resonance frequency and leakage inductance Lr and the excitation current for the primary winding P of the transformer Ta. The resonance current is provided, through a secondary winding S1 of the transformer Ta and an output rectifying diode D1, to an output capacitor Co and a load. When the resonance current passing through the secondary side becomes zero to leave only the excitation current, the switching element Q1 is turned off.
When the switching element Q1 is turned off, a voltage across the switching elements Q1 and Q2 has a quasi-voltage-resonance waveform based on a resultant value of the voltage resonance capacitor Crv and current resonance capacitor Cri and a resultant value of the inductance Lp of the primary winding P of the transformer Ta and the leakage inductance Lr. According to the relationship of Crv<<Cri, a resonance frequency at this time is also mostly determined by the voltage resonance capacitor Crv. The excitation current for the primary winding P passing through the switching element Q1 is translocated to a parasitic diode of the switching element Q2. After the voltage VQ2 across the switching element Q2 becomes zero, the switching element Q2 is turned on, to realize zero-voltage switching. Thereafter, the current IQ2 translocated to the switching element Q2 decreases, inverts its polarity, and passes through a MOSFET part of the switching element Q2. Thereafter, the above-mentioned operation is repeated.
The switching elements Q1 and Q2 each have a dead time and are alternately turned on/off. The switching element Q1 turns on with an ON width that may zero an ON-time resonance current, thereby substantially realizing zero-current switching. Namely, the ON time of the switching element Q1 is fixed and the ON time of the switching element Q2 is variable in order to adjust a charging voltage of the current resonance capacitor Cri and control the output voltage Vo.
The controller 1 illustrated in FIG. 3 is a PWM controller with the ON width of the low-side switching element Q1 being fixed and that of the high-side switching element Q2 being variable. On the other hand, a controller 1 illustrated in FIG. 6 is a PWM controller employing a fixed frequency. The frequency-fixed PWM controller of FIG. 6 employs a constant switching frequency and controls an ON/OFF ratio of the low- and high-side switching elements. In FIG. 6, the frequency-fixed PWM controller includes an oscillator PWOSC, an inverter INV, dead time generators DT1, and DT2, a level shifter LES, and buffers BUF1 and BUF2. If the ON width and resonance period of the low-side switching element are kept within proper ranges, the controller of FIG. 6 is usable like the controller of FIG. 3. The PWM controller of FIG. 6 is inexpensive. FIG. 7 illustrates waveforms at various locations of the PWM controller of FIG. 6. An example of this sort of frequency-fixed PWM controller is disclosed in Japanese Unexamined Patent Application Publication No. 2005-287257.
There are regulations and standards to restrict harmonic currents passing through commercial AC power source lines. For example, there is an international standard IEC61000-3-2. To minimize harmonic currents, an important factor is to bring an input current waveform closer to a sinusoidal wave. Generally, harmonic currents decrease if an input power factor is corrected. To correct the input power factor, the converter of the related art illustrated in FIG. 1 arranges the active filter circuit in front of the DC-DC converter part B. The active filter circuit is based on the step-up chopper to step up a lower voltage portion of a sinusoidal AC voltage and continuously pass an input current.